Method and apparatus for channel estimation for X-DSL communications

ABSTRACT

An apparatus and method is disclosed for channel estimation in an X-DSL communication device. The communication device may include physical or logical modems. The modems may implement one or more of a group of X-DSL protocols including: G.Lite, ADSL, VDSL, and HDSL. The apparatus may be used for determining the location and magnitude of discontinuities or faults within the communication medium to which the X-DSL communication device is coupled. The information provided by the device may be used for line qualification or repair. No additional equipment is required for channel estimation. Instead the apparatus may be located within a single modem or shared between a group of modems. An N bit pseudo random codeword injected into the transmit path is used to generate both a leakage signal and a plurality of reflected signals on the receive path. No timing information is needed from the transmit path. Instead a unique correlator is utilized on the receive path to extract timing information for the reflected signals relative to the leakage signal. The broad bandwidth of the codeword and its relatively long duration allow channel estimation at significantly higher signal-to-noise ratios and with greater degrees of accuracy than heretofore possible.

CROSS REFERENCE TO RELATED APPLICATION

This application is a continuation of copending U.S. application Ser.No. 09/757,036 filed Jan. 8, 2001 entitled “Method and Apparatus forChannel Estimation for X-DSL Communications” which claims the benefit ofprior filed Provisional Applications No. 60/175,012 filed on Jan. 7,2000 entitled “Ranging Algorithm for Channel Estimation for X-DSLApplication”. Each of the above-cited applications is incorporatedherein by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of Invention

The field of the present invention relates in general to X-DSLcommunication devices and more particularly to a method and apparatusfor channel estimation and fault detection in X-DSL communicationsystems.

2. Description of the Related Art

North American Integrated Service Digital Network (ISDN) Standard,defined by the American National Standard Institute (ANSI), regulatesthe protocol of information transmissions over telephone lines. Inparticular, the ISDN standard regulates the rate at which informationcan be transmitted and in what format. ISDN allows full duplex digitaltransmission of two 64 kilo bit per second data channels. These datarates may easily be achieved over the trunk lines, which connect thetelephone companies' central offices. The problem lies in passing thesesignals across the subscriber line between the central office and thebusiness or residential user. These lines were originally constructed tohandle voice traffic in the narrow band between 300 Hz to 3000 Hz atbandwidths equivalent to several kilo baud.

Digital Subscriber Lines (DSL) technology and improvements thereonincluding, G.Lite, ADSL, VDSL, HDSL all of which are broadly identifiedas X-DSL have been developed to increase the effective bandwidth ofexisting subscriber line connections, without requiring the installationof new fiber optic cable. An X-DSL modem operates at frequencies higherthan the voice band frequencies, thus an X-DSL modem may operatesimultaneously with a voice band modem or a telephone conversation.Currently there are over ten discrete X-DSL standards, including:G.Lite, ADSL, VDSL, SDSL, MDSL, RADSL, HDSL, etc.

One of the factors limiting the setup and operation of X-DSLcommunication systems is channel quality. Not all communicationsmediums, e.g. subscriber lines are capable of supporting various of theX-DSL protocols. On a subscriber line the presence of bridges, taps,isolators, filters etc. may effect channel quality to the point where agiven protocol may not be supported. Alternately, in operation channelquality may degrade due to improper repair or maintenance of thesubscriber line. In each instance the typical solution is to decouplethe subscriber line in the frame room of the PSTN central office and tocouple it to test equipment. The test equipment typically injects animpulse into the line and measures the amplitude and delay of each ofthe resultant echoes or reflections generated by the line. The pulsetypically has a duration shorter than the delay interval between any ofthe reflections in order to distinguish one reflection from the other.The energy injected into the line by the pulse determines the accuracyand completeness of the channel estimation produced thereby. A number offactors, however limit the energy of the pulse; i.e. its duration,subscriber line voltage/current limits, and the need to minimizeinterference with adjacent subscriber lines to which service is beingprovided.

What is needed are approaches to line estimation, qualification andfault detection that are lower in cost and which permit automation.

SUMMARY OF THE INVENTION

An apparatus and method is disclosed for channel estimation in an X-DSLcommunication device. The communication device may include physical orlogical modems. The modems may implement one or more of a group of X-DSLprotocols including: G.Lite, ADSL, VDSL, and HDSL. The apparatus may beused for determining the location and magnitude of discontinuities orfaults within the communication medium to which the X-DSL communicationdevice is coupled. The information provided by the device may be usedfor line qualification or repair. No additional equipment is requiredfor channel estimation. Instead the apparatus may be located within asingle modem or shared between a group of modems. An N bit pseudo randomcodeword injected into the transmit path is used to generate both aleakage signal and a plurality of reflected signals on the receive path.No timing information is needed from the transmit path. Instead a uniquecorrelator is utilized on the receive path to extract timing informationfor the reflected signals relative to the leakage signal. The broadbandwidth of the codeword and its relatively long duration allow channelestimation at significantly higher signal-to-noise ratios and withgreater degrees of accuracy than heretofore possible.

In an embodiment of the invention an apparatus for channel estimation ofa communication device with a transmit path and a receive path bothcoupled to a communication medium is disclosed. The apparatus includes agenerator, an analog-to-digital converter (“ADC”), and a correlator. Thegenerator couples to the transmit path for periodically injecting acodeword signal into the transmit path which effects both a leakagesignal on the receive path as well as reflected signals from variousportions of the communication medium. The ADC couples to the receivepath to digitize a composite received signal including both the leakagesignal and the reflected signals. The correlator correlates delaysbetween the leakage signal and each of the reflected signals to estimatechannel characteristics for the communication medium.

In an embodiment of the invention a method for channel estimation in acommunication device with a transmit path and a receive path bothcoupled to a communication medium is disclosed. The method comprisingthe acts of:

-   -   periodically injecting a codeword signal into the transmit path        which effects both a leakage signal on the receive path as well        as reflected signals from various portions of the communication        medium,    -   digitizing a composite received signal including both the        leakage signal and the reflected signals, and    -   correlating delays between the leakage signal and each of the        reflected signals to estimate channel characteristics for the        communication medium.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features and advantages of the present invention willbecome more apparent to those skilled in the art from the followingdetailed description in conjunction with the appended drawings in which

FIG. 1 shows an X-DSL communication system with a set of multi-modemulti-channel logical modems at a public switched telephone network(PSTN) central office (CO) coupled across a plurality of subscriberlines to a plurality of remote subscriber sites.

FIG. 2 is a detailed hardware block diagram of one of the modem linecards shown in FIG. 1.

FIG. 3 is a detailed logical block diagram showing basic logic blocksassociated with the apparatus for channel estimation of the currentinvention.

FIGS. 4AB show alternate embodiments of the correlator module shown inFIG. 3.

FIG. 5A is a graph showing an example of the codeword injected into thetransmit path of the in accordance with an embodiment of the invention.

FIGS. 5B–D show the leakage signal, a reflected signal, and thecomposite of the leakage and reflected signals respectively which areeffected on the receive path responsive to the injection of thecodeword.

FIG. 5E shows the correlation factors for the received signal.

FIG. 6 is a process flow diagram of the channel estimation processes.

DETAILED DESCRIPTION OF THE EMBODIMENTS

An apparatus and method is provided for minimizing in channel distortionin an X-DSL communication device is disclosed. The communication devicemay include physical or logical modems. The apparatus may beincorporated in an existing X-DSL architecture without additionalcircuitry. The apparatus may be used for determining the location andmagnitude of discontinuities or faults within the communication mediumto which the X-DSL communication device is coupled. The informationprovided by the device may be used for line qualification or repair. Noadditional equipment is required for channel estimation. Instead theapparatus may be located within a single modem or shared between a groupof modems. An N bit pseudo random codeword injected into the transmitpath is used to generate both a leakage signal and a plurality ofreflected signals on the receive path. The broad bandwidth of thecodeword and its relatively long duration allow channel estimation atsignificantly higher signal-to-noise ratios and with greater degrees ofaccuracy than heretofore possible. No timing information is needed fromthe transmit path. Instead a unique correlator is utilized on thereceive path to extract timing information for the reflected signalsrelative to the leakage signal. The apparatus may be applied with equaladvantage to communication protocols other than X-DSL. The apparatus maybe applied with equal advantage in wired and optical media.

FIG. 1 shows an X-DSL communication system with a set of multi-modemulti-channel logical modems at a public switched telephone network(PSTN) central office (CO) coupled across a plurality of subscriberlines to a plurality of remote subscriber sites. The system includes aCO 100 coupled via corresponding subscriber lines to remote sites150–154.

Each of the subscriber line connections terminates on the CO end, in theframe room 102 of the CO. From this room connections are made for eachsubscriber line via splitters and hybrids to both a DSLAM 104 and to thevoice band racks 106. The splitter shunts voice band communications todedicated line cards, e.g. line card 112 or to a voice band modem pool(not shown). The splitter shunts higher frequency X-DSL communicationson the subscriber line to a selected line card, e.g. line card 116,within DSLAM 104. The line cards of the current invention are universal,meaning they can handle any current or evolving standard of X-DSL andmay be upgraded on the fly to handle new standards.

Voice band call set up is controlled by a Telco switch matrix 114 suchas SS7. This makes point-to-point connections to other subscribers forvoice band communications across the public switched telephone network132. The X-DSL communications may be processed by a universal line cardsuch as line card 116. That line card includes a plurality of AFE's118–120 each capable of supporting a plurality of subscriber lines. TheAFEs are coupled via a packet based bus 122 to the DSP 124. Fordownstream communications the transmit path from the CO to the remotesite includes the DSP which modulates the data for each communicationchannel and the AFE which transforms the digital symbol packetsassembled by the DSP and converts them to an analog signal which isoutput on the subscriber line associated with the respective channel.For upstream communications the receive paths for each channel from theremote sites to the CO include conversion of each received channelwithin the corresponding AFE to a digitized data sample which is sent tothe DSP for demodulation. The DSP is capable of multi-protocol supportfor all subscriber lines to which the AFE's are coupled.

Communications between AFE's and DSP(s) may be packet based, in whichembodiment of the invention a distributed architecture such as will beset forth in the following FIG. 2 may be implemented. The line card 116is coupled to a back-plane bus 128 which may be capable of offloadingand transporting low latency X-DSL traffic between other DSPs for loadbalancing. The back-plane bus of the DSLAM also couples each line cardto the Internet 130 via server 108. Each of the DSLAM line cardsoperates under the control of a DSLAM controller 110 which handlesglobal provisioning, e.g. allocation of subscriber lines to AFE and DSPresources. The various components on the line card form a plurality oflogical modems each handling upstream and downstream communicationsacross corresponding subscriber lines. In an alternate embodiment of theinvention discrete modems would each couple to an associated one of thesubscriber lines rather than the logical modem shown. When an X-DSLcommunication is established on a subscriber line, a specific channelidentifier is allocated to that communication. That identifier is usedin the above mentioned packet based embodiment to track each packet asit moves in an upstream or downstream direction between the AFE and DSP.These modules, AFE and DSP, may be found on a single universal linecard, such as line card 116 in FIG. 2. They may alternately be displacedfrom one another on separate line cards linked by a DSP bus. In stillanother embodiment they may be found displaced from one another acrossan ATM network. There may be multiple DSP chipsets on a line card. In anembodiment of the invention the DSP and AFE chipsets may includestructures set forth in the figure for handling of multiple line codesand multiple channels.

A number of discontinuities 162–164 in the subscriber line 160 are shownfor example. These correspond with reflective elements of the line suchas repeaters, taps, isolators or breaks. Unique logic implemented inhardware or software on either or both the AFE or DSP is set forth inthe following FIGS. 2–6 for accurate channel estimation to locate andcharacterize each subscriber line including the discontinuities therein.

FIG. 2 is a detailed hardware block diagram of one of the modem linecards shown in FIG. 1. FIG. 2 shows a packet based multi-channeltransmission architecture within which the current invention may beimplemented. In this architecture a DSP 124 handles processing for anumber of channels of upstream and downstream subscriber linecommunications via a number of AFE's. Each AFE in turn accepts packetsassociated with a plurality of subscriber lines to which each AFE iscoupled. FIG. 2 shows a packet based raw data processing both between aDSP and AFE as well as within each DSP and AFE. Packet processingbetween DSP and AFE modules involves transfer of bus packets 294 eachwith a header portion 296 and data portion 298. The header containsinformation correlating the data with a specific channel and direction,e.g. upstream or downstream of communication. The data portion containsfor upstream traffic digitized samples of the received data for eachchannel and for downstream packets digitized symbols for the data to betransmitted on each channel. Packet processing within a DSP may involvedevice packets 286. The device packets may include a header 288, acontrol portion 290 and a data portion 292. The header serves toidentify the specific channel and direction. The header may containcontrol information for the channel to be processed. The control portion290 may also contain control portions for each specific component alongthe transmit or receive path to coordinate the processing of thepackets. Within the AFE the digitized data generated for the received(upstream data) will be packetized and transmitted to the DSP. Fordownstream data, the AFE will receive in each packet from the DSP thedigitized symbols for each channel which will be modulated in the AFEand transmitted over the corresponding subscriber line.

These modules, AFE and DSP, may be found on a single universal linecard, such as line card 116 in FIG. 1. They may alternately be displacedfrom one another on separate line cards linked by a DSP bus. In stillanother embodiment they may be found displaced across an ATM network.

DSP line card 116 includes one or more DSP's. In an embodiment of theinvention each may include structures set forth in the figure forhandling of multiple line codes and multiple channels. The line cardincludes, a DSP medium access control (MAC) 200 which handles packettransfers to and from the DSP bus 122. The MAC couples with a packetassembler/disassembler (PAD) 202. For received DSP bus packets, the PADhandles removal of the DSP bus packet header 296 and insertion of thedevice header 288 and control header 290 which is part of the devicepacket 286. The content of these headers is generated by the coreprocessor 212 using statistics gathered by the de-framer 222. Thesestatistics may include gain tables, or embedded operations channelcommunications from the subscriber side. The PAD embeds the requiredcommands generated by the core processor in the header or controlportions of the device packet header. Upstream device packets (Receivepackets) pass into a first-in-first-out FIFO buffer 208 which iscontrolled by FIFO controller 206. These packets correspond withmultiple protocols and multiple channels. Each is labeled accordingly.The receive processing engine 204 in this case a DMT engine fetchespackets and processes the data in them in a manner appropriate for theprotocol, channel and command instructions, if any, indicated by theheader. The processed data is then passed to the De-Frame and ReedSolomon Decoder 222. This module reads the next device packet andprocesses the data in it in accordance with the instructions orparameters in its header. The processed de-framed data is passed to thefinal FIFO buffer 226 which is controlled by controller 224. That datais then passed to the ATM pad 228 for wrapping with an ATM header andremoval of the device header. The ATM MAC 230 then places the data withan ATM packet on the ATM network 130 (see FIG. 1).

Control of the receive modules, e.g. DMT engine 204 and de-framerdecoder 222 as well as sub modules thereof is implemented as follows.The core processor 210 has DMA access to the FIFO buffer 226 from whichit gathers statistical information on each channel including gaintables, or gain table change requests from the subscriber as well asinstructions in the embedded operations portion of the channel. Thosetables 214 are stored by the core processor in memory 212. When a changein gain table for a particular channel is called for, the core processorsends instructions regarding the change in the header of the devicepacket for that channel via PAD 202 and writes the new gain table to amemory which can be accessed by the appropriate module, i.e. DMT module204, or the appropriate sub module thereof, as a packet corresponding tothat channel is received by the module. This technique of in bandsignaling with packet headers allows independent scheduling of actionson a channel by channel basis in a manner which does not require thedirect control of the core processor. Instead each module in thetransmit path can execute independently of the other at the appropriatetime whatever actions are required of it as dictated by the informationin the device header which it reads and executes.

This device architecture allows the DSP transmit and receive paths to befabricated as independent modules or sub modules which respond to packetheader control information for processing of successive packets withdifferent X-DSL protocols, e.g. a packet with ADSL sample data followedby a packet with VDSL sampled data. Within the DMT Rx engine 204 forexample, there may be sub modules with independent processing capabilitysuch as: a time domain equalizer, a cyclic prefix remover, a DFT, a gainscalar, a trellis decoder and a tone recorder, as well as filters, awindowers . . . etc. Each of these sub modules has its counterpart onthe DMT Tx engine 220 in the transmit path. Each of these mayindependently respond to successive device headers to change parametersbetween successive packets. For example as successive packets fromchannels implementing G.Lite, ADSL and VDSL pass through the DMT Txengine the number of tones will vary from 128 for G.lite, to 256 forADSL, to 2048 for VDSL. The framer and de-framer will use protocolspecific information associated with each of these channels to look fordifferent frame and super frame boundaries. The DMT receive engine 204implements processes for monitoring a monitor tone on the upstreamchannel during the setup and configuration phases of the method foradaptively minimizing out of band interference and in band distortion.The measured level of each tone is maintained by processor 210 in memory212. This same memory may be utilized for calculating the inversechannel model for each of the channels to determine the amount ofpre-distortion to be applied to downstream data on each of the channels.

On the downstream side, i.e. Transmit, the same architecture applies.ATM data which is unwrapped by PAD 228 is re-wrapped with a deviceheader the contents of which are again dictated by the core processor210. That processor embeds control information related to each channelin the packets corresponding to that channel. The device packets arethen passed to the FIFO buffer 232 which is controlled by controller234. The Framer and RS encoder 236 and or sub modules thereof thenprocesses these packets according to the information contained in theirheader or control portions of each device packet. The Framer thenupdates the device packet header and writes the resultant device packetto the DMT transmit module 220. This module accepts the data andprocesses it for transmission. Transmission processing may include: toneordering, trellis encoding, gain scaling, an IDFT, and cyclic prefixmodules each with independent ability to read and respond to deviceheaders. From the DMT Tx engine 220 each updated device packet with adigitized symbol(s) for a corresponding channel is placed in the FIFObuffer 216 under the control of controller 218. From this buffer thedevice packet is sent to PAD 202 where the device header is removed. TheDSP PAD places the DSP packet 294 with an appropriate header onto theDSP bus 122 for transmission to the appropriate AFE and the appropriatechannel and subscriber line within the AFE.

Because the data flow in the AFE allows a more linear treatment of eachchannel of information an out of band control process is utilized withinthe AFE. In contrast to the DSP device packets which are used tocoordinate various independent modules within the DSP the AFEaccomplishes channel and protocol changeovers with a slightly differentcontrol method.

A packet 294 on the bus 122 directed to AFE 120 is detected by AFE MAC240 on the basis of information contained in the packet header. Thepacket is passed to PAD 242 which removes the header 296 and sends it tothe core processor 244. The packet's header information includingchannel ID is stored in the core processor's memory 248. The informationis contained in a table 266. The raw data 298 is passed to a FIFO buffer252 under the control of controller 250. Each channel has a memorymapped location in that buffer.

On the transmit path, the interpolator 254 reads a fixed amount of datafrom each channel location in the FIFO buffer. The amount of data readvaries for each channel depending on the bandwidth of the channel. Theamount of data read during each bus interval is governed by entries inthe control table for each channel which is established during channelsetup and is stored in memory 248. The interpolator up samples the dataand low pass filters it to reduce the noise introduced by the DSP.Implementing interpolation in the AFE as opposed to the DSP has theadvantage of lowering the bandwidth requirements of the DSP bus 122.From the interpolator data is passed to the digital-to-analog converter(DAC) 260. The DAC converts the digitized symbol for each of the inputsignals on each of the input signal lines/channels to correspondinganalog signals. These analog signals are introduced to the amplificationstage 262, from which they are coupled to corresponding subscriberlines. The amplification stage is coupled to a power supply 266. Theparameters for each of the modules 254, 260, 262, i.e. filtercoefficients, amplifier gain etc. are controlled by the core processorusing control parameters stored during session set up. For example,where successive packets carry packets with G.Lite, ADSL, and VDSLprotocols the sample rate of the filter parameters for filter 254 andthe gain of the analog amplifiers within stage 262 will vary for eachpacket. This “on the fly” configurability allows a single transmit orreceive pipeline to be used for multiple concurrent protocols.

During line estimation or qualification the PRN generator 258 injects apseudo-random-noise into the transmit path of one or more of thesubscriber channels on the transmit path for transmission on thecorresponding subscriber channel. The injection of this codeword resultsin a composite signal on the receive path for the corresponding channelwhich includes a leakage signal resulting from the leakage between thetransmit and receive paths, i.e. self-NEXT, a.k.a. “Next EndCross-Talk”, as well as echoes resulting from the reflection of thetransmitted codeword of various discontinuities e.g. discontinuities162–164 within the corresponding subscriber line (see FIG. 1). Where thetransmit path includes filters (not shown) downstream of the location atwhich the codeword is injected, these filters may be switchablydecoupled from the transmit path during the transmission of thecodeword. The codeword may be injected periodically until the channel ischaracterized. In an embodiment of the invention the codeword has alength of pseudo randomness greater than the maximum delay timeassociated with the echo or reflections which result from thediscontinuities on the line. This code length maximizes the energyassociated with channel estimation by extending the injection intervalbeyond that associated with a single pulse.

On the upstream path, the receive path, individual subscriber linescouple to individual line amplifiers, e.g. 270–272, through splitter andhybrids (See FIG. 3.). Each channel is passed to dedicated ADC modules274–276. Next each channel may be subject to further filtering anddecimation 278. As discussed above in connection with the transmit path,each of these components is configured on the fly for each new packetdepending on the protocol associated with it. Each channel of data isthen placed in a memory mapped location of FIFO memory 282 under thecontrol of controller 280. Scheduled amounts of this data are moved toPAD 242 during each bus interval. The PAD wraps the raw data in a DSPheader with channel ID and other information which allows the receivingDSP to properly process it. In an alternate embodiment of the inventionthe same packet based control principal may be used in both the transmitand receive path to implement not only multiple protocols concurrentlybut alternate lines codes, e.g. CAP/QAM.

A correlator 284 is shown coupled to the receive path. The correlatorembodiments of which are set forth in the following FIGS. 4AB, operatesduring the generation of the codeword for channel estimation. Thecorrelator may be implemented as a discrete module, or as processesperformed on the processor, e.g. processor 244. The correlator operatesto receive from the corresponding ADC the digitized bits of thecomposite received signal which includes both the leakage signal and oneor more reflected signals from various discontinuities within thecorresponding subscriber line. The correlator generates an ordered setof correlation coefficients corresponding with various phasings of thecodeword with the composite signal. These coefficients may be stored inmemory in the correlator or in memory 248. The correlator detects peakswithin these ordered coefficients, identifies which of the peakscorresponds with the leakage signal, and from that informationdetermines the time delay, or offset, of each of the subsequentreflected signals along with their relative magnitudes. This informationis then used to characterize the associated subscriber linediscontinuities by location and by type using methods well known tothose skilled in the art. Channel estimation does not require the PRNgenerator 258 to couple with the codeword generator. Sincecommunications are packet based the actual time of transmission of thecodeword may be difficult to determine. Instead a timing reference isgenerated by the detection of the leakage peak within the receivedcomposite signal. Additionally, the channel estimator may be implementedon a single line card and may be switched between various channelsthereon. Finally, the length of the codeword results in a significantincrease in the amount of energy applied to the subscriber line duringchannel estimation which greatly improves the quality of the linecharacterization as opposed to prior art designs.

FIG. 3 is a detailed logical block diagram showing basic logic blocksassociated with the apparatus for channel estimation of the currentinvention. A single logical modem is shown with DSP 300 handling thedigital modulation/demodulation. These blocks may be implemented withinthe existing processing units of one or more physical or logical modems.A digital signal processor 300 is shown coupled to the transmit signalpath 302. During channel estimation a pseudo-random noise generator(PRN) 238 is coupled to the transmit signal path to periodically injecta codeword into the transmission path. The codeword is converted toanalog format in digital-to-analog converter (DAC) 260. Thecorresponding amplified signal is generated by amplifier 262 ondownstream signal line 304 which couples the transmit path to the hybrid306. The hybrid is shown coupled to the transformer 308 which in turn iscoupled to the subscriber line, e.g. 160 (See FIG. 1). The currentinvention may be applied with equal advantage in optical media as well.During channel estimation the transmission of each codeword effects aleakage signal corresponding with that codeword which is detected in theout-of-band upstream channel 312. This leakage signal and the varioussignals reflected from the discontinuities 162–164 are received on theupstream, received signal path, as a composite signal. Where theparticular X-DSL or other communication protocol includes filters oneither the transmit or receive path, e.g. filter 326 that filter maypreferably be switchably decoupled from the corresponding path so as notto limit channel estimation. The composite signal is amplified inamplifier 272 and the output is digitized within ADC 276. The digitizedoutput of the ADC is detected by correlator 284 which may be switchablycoupled with the receive path 314. The correlator generates correlationcoefficients, detects peaks therein including the leakage peak, andsequentially orders the peaks corresponding to each of the reflectedsignals from each of the discontinuities in the channel with respectthereto.

FIGS. 4AB show alternate embodiments of the correlator module shown inFIG. 3. In the first embodiment shown in FIG. 4A the correlatorincludes: a code word buffer, a received signal buffer 406, a multiplierbank, a summer 412 a memory 414 and a detector 418. The “N” bits of thecomplete codeword are stored within the individual locations, e.g.location 420 of codeword buffer 400. The codeword has a length whichcorresponds with the longest delay time for a reflected signal for theassociated channel. In the example shown the codeword P₀–P₁₅ is sixteenbits in length. A circular shifter 402 allows the individual bits of thewithin the code word register to be circular shifted during correlationof the composite signal on the receive path 314. A loader 404 coupled tothe input of the receive buffer 406 allows the composite received signalon line 314 the to be loaded into the receive buffer. The number of bitsin the received buffer is shown corresponding with the number of bits inthe codeword, though this need not be the case.

Once the code word and received signal are loaded into their respectivebuffers each bit of the codeword buffer is multiplied by a correspondingbit in the receive buffer by a corresponding multiplier within themultiplier bank and the results are placed in corresponding bitlocations within summer 412. Thus the codeword bit stored in bitlocation 420 in the codeword buffer is multiplied by multiplier 408times the codeword bit in location 430 in the receive buffer 406 and theoutput is placed in location 440 in summer 412. Then the contents of thesummer are added and the corresponding correlation coefficient K(L_(n))is stored in the coefficient table 462 within memory 414. Then thecodeword is shifted via shifter 402 and the process is repeated for thecalculation of the next correlation coefficient. This is continued atleast until the complete codeword has been rotated and the process maybe repeated is averaging of coefficients across multiple sample sets iscalled for as a way of increasing the accuracy of the result.

The memory also includes program code for effecting the processes shownin the following FIG. 6. After the coefficient table 462 is completedwith an ordered set of the calculated coefficients one or moreadditional runs of the aforementioned logic may be accomplished in whichcase the contents of the coefficient table are averaged in averager 416to produce a single coefficient table. In either case, the orderedcorrelation coefficients of table 462 are provided to detector 418.Detector 418 includes the peak detector 450, a leakage peak detector452, and the sequencer 454. The peak detector 450 determines thelocation and number of the peaks within the ordered coefficient list.This may be done using a number of techniques well known to thoseskilled in the art including a fixed threshold a crossing and recrossingof which signifies a peak or valley. Next the leakage peak detector 452determines on the basis of the relative spacing between peaks whichamong the peaks is the leakage peak. The leakage peak in an embodimentof the invention is the first peak following the tail of the echoes. Itis the peak the location of which is spaced furthest apart in thecorrelation coefficient table from a preceding peak. The sequencer thenorders each of the reflective peaks which correspond to each of thereflective signals with respect to leakage peak in terms of both theiroffset in relative magnitude. This information is then provided onoutput 328. This information may then be processed further to determinethe location of the discontinuity based on signal propagation timeswithin the subscriber line and the offset of the correspondingreflection peak from the leakage peak. Additionally the magnitude andother parameters of each reflective peak may be compared with knownvalues for various types of line discontinuities such as taps, andbridges to identify the discontinuity not only by location but by type.Channel or line estimation may also be used to disqualify, or qualify aparticular line for one or more of the X-DSL protocols. A line with manydiscontinuities might not qualify for VDSL service but may qualify forG.Lite or other low bandwidth ones of the X-DSL protocols.

In FIG. 4B an alternate embodiment of the correlator 284 is shown. Thisembodiment is identical in all respects to the embodiment shown in FIG.4A with the exception that the individual bits within the code wordbuffer 400 are fixed and the bits within the receive buffer 404 areshifted via shifter 462 as each new bit of the composite signal isreceived on signal line 314.

In alternate embodiments of the invention the multipliers of themultiplier bank may be replaced with simpler circuitry where thecodeword is limited to a random sequence of “0”s together with “1”s. Ina first of these embodiments where the incoming data is in sign plusmagnitude format the multiplier can be replaced by a simple circuitusing an “XOR” gate. Alternately, where the codeword bits are expressedin twos complement format the multiplier may be replaced by amultiplexer coupled to the corresponding bit of the codeword buffer anda pair of inputs, one of which is inverted, coupled to the correspondingbit of the receive buffer.

In alternate embodiments of the invention the logic shown in FIGS. 4A–Bmay be implemented in hardware, or software on a physical modem, in theDSP or AFE of a logical modem, by software executed on a processor or bydedicated or shared circuitry.

FIG. 5A is a graph showing an example of the codeword injected into thetransmit path in accordance with an embodiment of the invention. Thecodeword is 16 bits in length and includes a pseudo random sequence of+/− “1”s commencing with bit P₀ referenced as 500 shown in a dottedpattern. This signal may be periodically injected by the PRN generatorinto a corresponding subscriber line transmit path for channelcharacterization.

FIGS. 5B–D show the leakage signal commencing at 502, a reflected signalcommencing at 504, and the composite signal resulting from the leakageand reflected signals respectively which are effected on the receivepath responsive to the injection of the codeword shown in FIG. 5A intothe transmit path.

FIG. 5E shows the correlation factors for the received signal. Two peaksin the ordered set of coefficients are shown, peak 530 and peak 532.Depending on the initial sampling of the composite received signal thesecould occur in the order shown or could be shifted across/around thecorrelation graph. In the example shown the leakage peak 530 follows theinterval of greatest separation between peaks, i.e. interval 520. Thiscorresponds with the tail of the reflections from the furthest end ofthe subscriber line in which any reflections are most severelyattenuated. The leakage peak is separated by interval 522 from the firstcorrelation coefficient peak 532 which peak corresponds with the firstreflected signal from the first discontinuity on the correspondingsubscriber line, e.g. line 160, discontinuity 162 (See FIG. 1).

The following is a mathematical exposition of an embodiment of theprocesses associated with the correlator of the current invention. Thechannel estimation and/or line qualification utilizes, in thisembodiment of the invention, a pseudo-random sequence to detect thediscontinues and bridge taps in the line. The pseudo-random sequencescan be generated in various ways, but they all possess the followingproperty. $\begin{matrix}{{\sum\limits_{i}\;{{P\left( {n - q_{j}} \right)}{P\left( {n - l_{i}} \right)}}} = \left\{ \begin{matrix}0 & {q_{i} = l_{i}} \\0 & {q_{j} \neq l_{j}}\end{matrix} \right.} & {{Equation}\mspace{14mu} 1}\end{matrix}$where delay is defined as a circular shift. The above property statesthat the dot product of a sequence with its circularly shifted versionis substantially zero and equal to a non-zero positive constant for zeroshift case. One of the ways to produce a pseudo-random sequence is bymeans of feedback shift-register (m-sequence) in which case equation (1)is expressed as: $\begin{matrix}{{\sum\limits_{n = 0}^{N - 1}\;{{P\left( {n - q_{j}} \right)}{P\left( {n - l_{j}} \right)}}} = \left\{ \begin{matrix}N & {q_{j} = l_{i}} \\{- 1} & {q_{j} \neq l_{j}}\end{matrix} \right.} & {{Equation}\mspace{14mu} 2}\end{matrix}$where N is the period of the pseudo-random sequence.

The system starts by transmitting a pre-defined pseudo-random sequencegenerated in time. The duration of the transmitted pre-definedpseudo-random sequence should be longer than multiple periods of thesequence. The received signal is the superposition of the leakage of thetransmitted signal via the hybrid 308 (See FIG. 3) and the reflectionsof the signal due to discontinuities in the line. The received signal ismathematically modeled as $\begin{matrix}{{r(n)} = {\sum\limits_{i = 0}^{M}\;{\alpha_{1}{p\left( {n - \tau_{i}} \right)}}}} & {{Equation}\mspace{14mu} 3}\end{matrix}$where α₀p(n−τ₀) is the leaked transmit signal via the hybrid circuitry,and the terms corresponding to i=1 . . . M is caused by the reflectionof the discontinues of the line. At the receiver, the received signalr(n) is cross-correlated against the same pseudo-random sequencegenerated locally. Assuming a feedback shift-register m-sequence wasutilized as the pseudo-random sequence, the cross-correlation functionis presented below. $\begin{matrix}{{{{K(l)} = {{\sum\limits_{n = 0}^{N -}\;{{r(n)}{p\left( {n - l} \right)}}} = {\left( {\sum\limits_{i = 0}^{M}\;{\left( {N + 1} \right){{\alpha\delta}\left( {i - \tau_{i}} \right)}}} \right) - 1}}}{{{{for}\mspace{20mu} l} = 0},\ldots\mspace{14mu},{N - {1\mspace{14mu}{where}}}}}\mspace{34mu}} & {{Equation}\mspace{14mu} 4} \\{{\delta\left( {l - k} \right)} = \left\{ \begin{matrix}1 & {k = l} \\0 & {k \neq l}\end{matrix} \right.} & {{Equation}\mspace{14mu} 5}\end{matrix}$In the above calculation of K(l), it is assumed that p(n) is periodicwith period N. The receiver by observing K(l) will utilize thecalculated the values of α_(i) for i=1 . . . M and d_(i)=τ_(i)−τ₀ fori=1, . . . M to estimate the line insertion loss. Moreover, thecross-correlation function K(l) can also be calculated as follow$\begin{matrix}{{{K(l)} = {{\sum\limits_{n = 0}^{N - 1}\;{{r\left( {n - l} \right)}{p(n)}}} = {\left( {\sum\limits_{i = 0}^{M}\;{\left( {N + 1} \right)\alpha_{i}{\delta\left( {l - \tau_{i}} \right)}}} \right) - 1}}}{{{{for}\mspace{20mu} l} = 0},\ldots\mspace{14mu},{N - 1}}} & {{Equation}\mspace{14mu} 6}\end{matrix}$Assuming the presence of noise in the receive signal, estimating K(l)using Eq. 6 has the advantage of being able to average the receivedsignal overtime to reduce the noise effect. The averaging operation isshown below $\begin{matrix}{{{\overset{\_}{K}(l)} = {{\frac{1}{P}{\sum\limits_{i = 0}^{P}\;{{K\left( {l + {Ni}} \right)}\mspace{14mu}{for}\mspace{20mu} l}}} = 0}},\ldots\mspace{14mu},{N - 1}} & {{Equation}\mspace{14mu} 7}\end{matrix}$The line insertion loss will be an indication of the possible datathroughput of the line, which will determine if the line is qualified tocarry a given DSL service. FIG. 5A shows one period of m-sequence, ofperiod 2⁴−1=15. The transmitted sequence is a repeated version of them-sequence shown in FIG. 5A. The received sequence shown in FIG. 5D, isa super-position of the transmit leakage via hybrid shown in FIG. 5B aswell as the reflection shown in FIG. 5C. The shift 522 between the twois τ_(l)−τ₀=3. The relative magnitudes are α_(i)=0.5, α_(i)=0.25respectively. The cross correlation shown in FIG. 5E is done in acircular fashion.

FIG. 6 is a process flow diagram of the channel estimation processescorresponding with the embodiment of the correlator shown in FIG. 4A.Processing begin at start block 600 from which control is passed toprocess 602. In process 602 any interfering transmit filters in thetransmit or receive path are de-coupled from the corresponding path.Control is then passed to process 604. In process 604 the PRN generator238 (See FIG. 380) injects a code word with period “N” into the transmitpath. Control then passes to process 606. In process 606 a receivesignal of duration “N” bits is loaded into the receive buffer 406 (SeeFIG. 4A). Control then passes to process 608 in which the N bits of thePRN codeword are loaded into the corresponding locations within codewordbuffer 400 (See FIG. 4A). Control then passes to process 610. In process610 to the first/next shift of the PRN codeword in buffer 400 isaccomplished by shifter 402 (See FIG. 4A). Control than passes toprocess 612. In process 612 each corresponding bit within the codewordbuffer is multiplied by the corresponding bit in the receive samplebuffer 406 and passed to the corresponding locations in summer 412 (SeeFIG. 4A). Control then passes to process 614. In process 614 themultiplication results are summed to produce the correspondingcorrelation coefficient K(L_(n)). Next in process 616 this coefficientis stored in the coefficient table 462 (See FIG. 4A). Control thanpasses decision block 618. In decision block 618 a determination is madeas to whether there are any remaining shifts of the codeword buffer. Ifthere are then control returns to process 610. If not control passes todecision process 620. In decision block 620 a determination is made asto whether another receive sequence will be processed, in order tofurther improve the accuracy of the set of correlation coefficientsthrough averaging thereof across multiple correlation sets. If so,control returns to process 606. Alternately control passes to process622.

In process 622 the ordered set of correlation coefficients is retrievedfrom memory 414 and the peaks within the correlation coefficients aredetermined. Control than passes to process 624 for a determination ofthe spacing between peaks including the spacing between the last peakand the first peak. Next in process 626 the leakage peak is determinedon the basis of the differences determined in process 624. The leakagepeak as discussed above follows the greatest inter peakseparation/spacing. The leakage peak follows the tail of the reflectedsignals which corresponds with this spacing. Depending on the phasing ofthe sampling of the composite signal on the receive signal path theleakage peak may correspond with the first, the last, or an intermediateone of the peaks within the ordered set of correlation coefficients.Control than passes to process 628 in which the amplitude and offset ofeach subsequent peak with respect to the leakage peak determined. Theseresults may be subject to further processing for the determination ofline characteristics i.e. discontinuity location and discontinuity type.

The processes shown in FIG. 6 may be altered to correspond with theembodiment of the correlator shown in FIG. 4B by shifting the receivebuffer rather than the codeword buffer.

The foregoing description of a preferred embodiment of the invention hasbeen presented for purposes of illustration and description. It is notintended to be exhaustive or to limit the invention to the precise formsdisclosed. Obviously many modifications and variations will be apparentto practitioners skilled in this art. It is intended that the scope ofthe invention be defined by the following claims and their equivalents.

1. An apparatus for channel estimation in a communication device havinga transmit path and a receive path both coupled to a communicationmedium, and the apparatus comprising: a pseudo-random noise generator(PRN) coupled to the transmit path to inject a codeword consisting of‘+1’s together with ‘−1’s or ‘0’s together with ‘1’s into the transmitpath; a correlator coupled to the receive path to generate an orderedset of correlation coefficients corresponding with successive phasingsof the codeword with respect to a received signal, and the correlatorincluding: a detector to detect peaks within the ordered set ofcorrelation coefficients including both a peak corresponding with aleakage signal together with at least one other peak corresponding to areflection of the injected codeword by the communication medium, and thedetector determining at least one of the offset between peaks or arelative magnitude of the peaks, thereby estimating the channelcharacteristics across the communication medium; a plurality of XORgates each with an output and a pair of inputs a first of which pair ofinputs couples to a corresponding bit of the codeword and a second ofwhich pair of inputs couples to a corresponding sample of the receivedsignal; a shifter to shift the codeword with respect to thecorresponding samples of the received signal or vice-versa; and a summercoupled to the outputs of each bit of the plurality of XOR gates to sumthe outputs of the plurality of the plurality of XOR gates codeword oneach shift of the shifter, thereby generating successive ones of theordered set of correlation coefficients.
 2. An apparatus for determiningthe location and magnitude of discontinuities or faults within thecommunication medium to which the X-DSL communication device is coupled.The information provided by the divice may be used for linequalification or repair. No additional equipment is required for channelestimation. Instead the apparatus may be located within a single modemor shared between a group of modems. An N bit pseudo random codewordinjected into the transmit path is used to generate both a leakagesignal and a plurality of reflected signals on the receive path. Notiming information is needed from the transmit path. Instead a uniquecorrelator is utilized on the receive path to extract timing informationfor the reflected signals relative to the leakage signal. The broadbandwidth of the codeword and its relatively long duration allow channelestimation at significantl higher signal-to-noise ratios and withgreater degrees of accuracy than heretofore possible.
 3. A method forchannel estimation in a communication device having a transmit path anda receive path both coupled to a communication medium, and the methodcomprising the acts of: injecting a pseudo-random codeword into thetransmit path of the communication device; generating from the receivepath an ordered set of correlation coefficients corresponding withsuccessive phasings of the pseudo-random codeword with respect to areceived signal resulting from the injecting act; detecting peaks withinthe ordered set of correlation coefficients including both a peakcorresponding with a leakage signal together with at least one otherpeak corresponding to a reflection of the injected pseudo-randomcodeword by the communication medium; determining which among the peaksdetected detected in the detecting act corresponds with the leakagepeak; and sequentially ordering the peaks corresponding with a time ofreceipt of each of the reflected signals with respect to the leakagesignal to estimate channel characteristics for the communication medium;and determining at least one of the offset between peaks or a relativemagnitude of the peaks, thereby estimating the channel characteristicsacross the communication medium.